This invention relates to wideband code division multiple access (WCDMA) for a communication system and more particularly to modulation of primary or secondary synchronization codes to indicate space-time transmit diversity for WCDMA signals.
Present code division multiple access (CDMA) systems are characterized by simultaneous transmission of different data signals over a common channel by assigning each signal a unique code. This unique code is matched with a code of a selected receiver to determine the proper recipient of a data signal. These different data signals arrive at the receiver via multiple paths due to ground clutter and unpredictable signal reflection. Additive effects of these multiple data signals at the receiver may result in significant fading or variation in received signal strength. In general, this fading due to multiple data paths may be diminished by spreading the transmitted energy over a wide bandwidth. This wide bandwidth results in greatly reduced fading compared to narrow band transmission modes such as frequency division multiple access (FDMA) or time division multiple access (TDMA).
New standards are continually emerging for next generation wideband code division multiple access (WCDMA) communication systems as described in Provisional U.S. patent application Ser. No. 60/082,671,filed Apr. 22, 1998,and incorporated herein by reference. These WCDMA systems are coherent communications systems with pilot symbol assisted channel estimation schemes. These pilot symbols are transmitted as quadrature phase shift keyed (QPSK) known data in predetermined time frames to any receivers within range. The frames may propagate in a discontinuous transmission (DTX) mode. For voice traffic, transmission of user data occurs when the user speaks, but no data symbol transmission occurs when the user is silent. Similarly for packet data, the user data may be transmitted only when packets are ready to be sent. The frames are subdivided into fifteen equal time slots of 0.67 milliseconds each. Each time slot is further subdivided into equal symbol times. At a data rate of 30 KSPS, for example, each time slot includes twenty symbol times. Each frame includes pilot symbols as well as other control symbols such as transmit power control (TPC) symbols and rate information (RI) symbols. These control symbols include multiple bits otherwise known as chips to distinguish them from data bits. The chip transmission time (TC), therefore, is equal to the symbol time rate (T) divided by the number of chips in the symbol (N).
Previous studies have shown that multiple transmit antennas may improve reception by increasing transmit diversity for narrow band communication systems. In their paper New Detection Schemes for Transmit Diversity with no Channel Estimation, Tarokh et al. describe such a transmit diversity scheme for a TDMA system. The same concept is described in A Simile Transmitter Diversity Technique for Wireless Communications by Alamouti. Tarokh et al. and Alamouti, however, fail to teach such a transmit diversity scheme for a WCDMA communication system.
Referring to FIG. 1, there is a simplified block diagram of a typical transmitter using Space-Time Transit Diversity (STTD) of the prior art. The transmitter circuit receives pilot symbols, TPC symbols, RI symbols and data symbols on leads 100, 102, 104 and 106, respectively. Each of the symbols is encoded by a respective STTD encoder. Each STTD encoder produces two output signals that are applied to multiplex circuit 120. The multiplex circuit 120 produces each encoded symbol in a respective symbol time of a frame. Thus, a serial sequence of symbols in each frame is simultaneously applied to each respective multiplier circuit 124 and 126. A channel orthogonal code Cm is multiplied by each symbol to provide a unique signal for a designated receiver. The STTD encoded frames are then applied to antennas 128 and 130 for transmission.
Turning now to FIG. 2, there is a block diagram showing signal flow in an STTD encoder of the prior art that may be used with the transmitter of FIG. 1 for pilot symbol encoding. The pilot symbols are predetermined control signals that may be used for channel estimation and other functions. The encoding pattern of STTD encoder 112 is given in TABLE I. The STTD encoder receives pilot symbol 11 at symbol time T, pilot symbol S1 at symbol time 2T, pilot symbol 11 at symbol time 3T and pilot symbol S2 at symbol time 4T on lead 100 for each of sixteen time slots of a frame. The encoder has an exemplary data rate of 32 KSPS and produces a sequence of four pilot symbols for each of two antennas corresponding to leads 204 and 206, respectively, for each of the sixteen time slots of TABLE I. The STTD encoder produces pilot symbols B1,S1,B2 and S2 at symbol times T-4T, respectively, for a first antenna at lead 204. The STTD encoder simultaneously produces pilot symbols B1, xe2x88x92S2*, xe2x88x92B2 and S1* at symbol times T-4T, respectively, at lead 206 for a second antenna. Each symbol includes two bits representing a real and imaginary component. An asterisk indicates a complex conjugate operation or sign change of the imaginary part of the symbol. Pilot symbol values for the first time slot for the first antenna at lead 204, therefore, are 11, 11, 11 and 11. Corresponding pilot symbols for the second antenna at lead 206 are 11, 01, 00 and 10.
The bit signals rj(i+xcfx80j) of these symbols are transmitted serially along respective paths 208 and 210. Each bit signal of a respective symbol is subsequently received at a remote mobile antenna 212 after a transmit time xcfx80 corresponding to the jth path. The signals propagate to a despreader input circuit (not shown) where they are summed over each respective symbol time to produce input signals Rj1, Rj2, Rj3 and Rj4 corresponding to the four pilot symbol time slots and the jth of L multiple signal paths as previously described.
The input singals corresponding to the pilot symbols for each time slot are given in equations [5-8]. Noise terms are omitted for simplicity. Received signal Rj1 is produced by pilot symbols (B1,B1) having a constant value (11,11) at symbol time T for all time slots. Thus, the received signal is equal to the sum of respective Rayleigh fading parameters corresponding to the first and second antennas. Likewise, received signal Rj3 is produced by pilot symbols (B2,xe2x88x92B2)having a constant value (11,00) at symbol time 3T for all time slots. Channel estimates for the Rayleigh fading parameters corresponding to the first and second antennas, therefore, are readily obtained from input signals Rj1 and Rj3 as in equations [9] and [10].
Rj1=xcex1j1+xcex1j2xe2x80x83xe2x80x83[5]
Rj2=xcex1j1S1xe2x88x92xcex1j2S2*xe2x80x83xe2x80x83[6]
Rj3=xcex1j1xe2x88x92xcex1j2xe2x80x83xe2x80x83[7]
Rj4=xcex1j1S1+xcex1j2S1*xe2x80x83xe2x80x83[8]
xcex1j1=(Rj1+Rj3)/2xe2x80x83xe2x80x83[9]
xcex1j2=(Rj1xe2x88x92Rj3)/2xe2x80x83xe2x80x83[10]
Referring now to FIG. 3, there is a schematic diagram of a phase correction circuit of the prior art that may be used with a remote mobile receiver. This phase correction circuit receives input signals Rj1 and Rj2 on leads 324 and 326 at symbol times 2T and 4T, respectively. Each input signal has a value determined by the transmitted pilot symbols as shown in equations [6] and [8], respectively. The phase correction circuit receives a complex conjugate of a channel estimate of a Rayleigh fading parameter xcex1j1* corresponding to the first antenna on lead 302 and a channel estimate of another Rayleigh fading parameter xcex1j2 corresponding to the second antenna on lead 306. Complex conjugates of the input signals are produced by circuits 308 and 330 at leads 310 and 322, respectively. These input signals and their complex conjugates are multiplied by Rayleigh fading parameter estimate signals and summed as indicated to produce path-specific first and second symbol estimates at respective output leads 318 and 322 as in equations [11] and [12].
Rj1xcex1j1*+Rj2*xcex1j2=(|xcex1j1|2+|xcex1j2|2)S1xe2x80x83xe2x80x83[11]
xe2x88x92Rj1*xcex1j2+Rj2xcex1j1*=(|xcex1j1|2+|xcex1j2)S2xe2x80x83xe2x80x83[12]
These path-specific symbol estimates are then applied to a rake combiner circuit 404 (FIG. 4) to sum individual path-specific symbol estimates, thereby providing net soft symbols or pilot symbol signals as in equations [13] and [14].                                           S            ~                    1                =                                            ∑                              j                =                1                            L                        ⁢                                          R                j                1                            ⁢                              α                j                                  1                  *                                                              +                                    R              j                              2                *                                      ⁢                          α              j              2                                                          [        13        ]                                                      S            ~                    2                =                                            ∑                              j                =                1                            L                        ⁢                                          -                                  R                  j                                      1                    *                                                              ⁢                              α                j                2                                              +                                    R              j              2                        ⁢                          α              j                              1                *                                                                        [        14        ]            
These soft symbols or estimates provide a path diversity L and a transmit diversity 2. Thus, the total diversity of the STTD system is 2L. This increased diversity is highly advantageous in providing a reduced bit error rate.
Referring now to FIG. 4, there is a simplified diagram of a mobile communication system that may use the phase correction circuit (FIG. 3). The mobile communication system includes an antenna 400 for transmitting and receiving external signals. The diplexer 402 controls the transmit and receive function of the antenna. Multiple fingers of rake combiner circuit 404 combine received signals from multiple paths. Symbols from the rake combiner circuit 404,including pilot symbol signals of equations [13] and [14], are applied to a bit error rate (BER) circuit 410 and to a Viterbi decoder 406. Decoded symbols from the Viterbi decoder are applied to a frame error rate (FER) circuit 408. Averaging circuit 412 produces one of a FER and BER. This selected error rate is compared to a corresponding target error rate from reference circuit 414 by comparator circuit 416. The compared result is applied to bias circuit 420 via circuit 418 for generating a signal-to-interference ratio (SIR) reference signal on lead 424.
Pilot symbols from the rake combiner 404 are applied to the SIR measurement circuit 432. These pilot symbols are obtained from a common pilot channel similar to a broadcast channel. The SIR measurement circuit produces a received signal strength indicator (RSSI) estimate from an average of received pilot symbols. The SIR measurement circuit also produces an interference signal strength indicator (ISSI) estimate from an average of interference signals from base stations and other mobile systems over many time slots. The SIR measurement circuit produces an SIR estimate from a ratio of the RSSI signal to the ISSI signal. This SIR estimate is compared with a target SIR by circuit 426. This comparison result is applied to TPC command circuit 430 via circuit 428. The TPC command circuit 430 sets a TPC symbol control signal that is transmitted to a remote base station. This TPC symbol instructs the base station to either increase or decrease transmit power by preferably 1 dB for subsequent transmission.
Turning now to FIG. 5, there is a diagram showing a weighted multi-slot averaging (WMSA) circuit 732 of the prior art for channel estimation. In operation, a signal buffer circuit 706FIG. 7) receives individual frames of data having a predetermined time period of 10 milliseconds. Each frame of the PCCPCH is subdivided into sixteen equal time slots of 0.625 milliseconds each. Each time slot, for example time slot 528, includes a respective set of pilot symbols 520 and data symbols 529. The WMSA circuit (FIG. 5) samples pilot symbols from preferably 6 time slots for a Doppler frequency of less than 80 Hz and from preferably 4 time slots for a Doppler frequency of 80 Hz or more. These sampled pilot symbols are multiplied by respective weighting coefficients xcex11 through xcex1N and combined by the adder circuit xe2x80x9c+xe2x80x9d to produce a channel estimate. This channel estimate is used to correct the phase of received data symbols in time slot 527 estimate for a respective transmit antenna.
Referring now to FIG. 6, there is a despreader circuit of the prior art. Received signals from mobile antenna 212 propagate to the despreader circuit where they are summed over each respective symbol time to produce output signals Rj1 and Rj2 corresponding to the jth of L multiple signal paths as previously described. The despreader circuit receives the ith of N chip signals per symbol together with noise along the jth of L multiple signal paths at a time xcfx80j after transmission. Both here and in the following text, noise terms are omitted for simplicity. This received signal rj(i+xcfx80j) at lead 600 is multiplied by a channel orthogonal code signal Cm(i+xcfx80j) at lead 604 that is unique to the receiver. Each chip signal is summed over a respective symbol time by circuit 608 and produced as first and second output signals Rj1 and Rj2 on leads 612 and 614 as in equations [1-2], respectively. Delay circuit 610 provides a one-symbol delay T so that the output signals are produced simultaneously.
This arrangement advantageously provides additional gain at the mobile communication system by multiple path transmit antenna diversity from a remote base station. The mobile unit, however, must be compatible with base stations having a single transmit antenna as well as base stations having transmit antenna diversity. A problem arises, therefore, when the mobile communication system is initially powered up or when it passes from one cell to another cell. The mobile unit must not only determine which of several base signals offers a preferable signal strength. It must also determine whether the base station offers transmit antenna diversity. If the mobile unit incorrectly decodes a received signal and assumes no transmit diversity, it loses the improved gain of transmit diversity. Alternatively, if the mobile unit incorrectly decodes a received signal and assumes transmit diversity, multiple fingers of the rake combiner circuit 404 contribute noise to the received signal.
A previous method of blind diversity detection by a mobile unit was presented in U.S. patent application Ser. No. 09/373,855, filed Aug. 13, 1999, and incorporated herein by reference. Therein, a method was disclosed to detect diversity transmission at a base station based on received signal strength. A problem with this method of blind diversity detection arises when a mobile unit must decode pilot symbols from broadcast channels (BCCH) of multiple base stations. This detection may require 250 milliseconds for each base station. Alternatively, if the base station communicates its diversity status to the mobile unit through a third layer (L3) message, this message still requires Viterbi decoding. Thus, previous methods of diversity detection require time and power to detect diversity and thereby optimally select a base station during power-up or during a soft handoff.
The foregoing problems are resolved by a circuit for detecting a transmit diversity signal, comprising a first circuit (706) arranged to receive a first synchronization code. The first synchronization code is modulated by a data signal. The first circuit produces a first output signal. A second circuit (732) is arranged to receive a plurality of predetermined signals. The second circuit produces a channel estimate. A detection circuit (710, 712) is arranged to receive the first output signal and the channel estimate. The detection circuit produces a signal corresponding to the data signal.
The present invention reduces transmit diversity detection time. Diversity detection is accomplished without Viterbi decoding.